The preamplifier stage of the unit is based on IC1 which is a low noise op amp having a FET input stage. This is used in the conventional inverting audio amplifier mode and the negative feedback network, R1, 4, sets the voltage gain at about 40dB. (100 times C11 reduces the gain slightly at high frequencies in order to obtain an improved signal to noise ratio.

C4 couples the output from the preamplifier to volume control, RV1, and from here the signal is coupled to the power amplifier by C5. The output stage uses the TBA820M, a class B amplifier which will give an output power of a few hundred milliwatts rms. The closed loop voltage gain of the device is determined by the value of R6, about 25dB. (180 times) with the specified value. This gives the required very high overall gain in conjunction with the preamplifier’s gain. C7, R7 and C8 are needed in order to maintain stability.

The quiescent current consumption of the unit is only about 5mA, but this rises to as much as 50mA or so at high volume levels. The best position for the pick-up coil on the telephone base (not the handset) can be located with a little experimentation.

R1 and R2 are used to bias the unit to give the optimum quiescent output potential and they provide overall negative feedback, which improves the quality of reproduction. D1 and C4 are boot strapping components, enabling the gate drive voltage to Q3 to go above the positive supply potential, giving improved efficiency to the circuit. R3 is the main collector load for Q2 and PR1 is used to give a standing bias on the output transistors that gives a quiescent current consumption of about 25mA. The thermal compensation circuitry normally used is totally unnecessary in this circuit, since VMOS devices do not suffer from thermal runaway. In fact the quiescent bias current will drop slightly as the output devices heat up, but not sufficiently to give rise to significant crossover distortion.

C2 and C5 provide DC blocking at the input and output respectively, while C1 is a supply decoupling component. C3 gives a degree of high frequency attenuation and aids the stability of the circuit.

Although the current in the driver stage, only about 1mA, may seem to be totally inadequate, it is in fact more than sufficient since the VMOS devices have extremely high input impedances and consume no significant, input current. This is one of their main advantages over bipolar devices. One disadvantage in this particular application is lower efficiency due to the higher threshold voltages and on resistance of VMOS transistors in comparison to bipolar devices. However, the circuit will give an output of 10W rms using a supply voltrage of about 33V or so (with a current drain of up to about 600mA). An input of about 500mV rms is needed for maximum output.

Note: The output devices do not have internal zener protection diodes and the appropriate handling precautions should be taken.

The output signal level from a cassette tape head is typically about 500 micro Volt or so at middle audio frequencies for a mono head and about half this level for a stereo type. The preamplifier must, therefore, provide a considerable amount of voltage gain in order to match this to a hi-fi amplifier, since these require a signal level about 1,000 times higher. It is also necessary for the preamplifier to provide equalization, because the output from a tape head rises at a rate of 6dB per octave. However at higher audio frequencies, tape heads are not very efficient and require a much less rolloff.

Q1 and Q2 are used in a conventional two stage, direct coupled, common emitter amplifier and the frequency-selective negative feed-back through C3 and R4 provides the appropriate equalization. These also set the midband voltage gain of the input stage at about 46dB. With such a low input level it is obviously necessary to use low noise transistors (Such as the BC109C) in order to obtain good results. Running Q1 at a low collector Current, about 200uA, also helps.

Q3 is used as a low gain common emitter stage, which provides the additional amplification. R9 introduces negative feedback, which controls the voltage gain of Q3 and the specified value gives a gain of about 14dB. For a stereo unit R9 should be reduced to 390R in order to give increased gain, to compensate for the lower output of a stereo tape head.

When playing a Dolby B encoded cassette SVV1 can be closed; this gives a small degree of treble cut which provides a reasonably flat overall response.

It is a conventional second order filter circuit having passive high pass filter formed by the series capacitance C3 and C4, plus the parallel resistance of R2 and R3 (the latter also being used to bias emitter follower transistor Q1). A passive filter of this type gives only a very slow initial roll off, and an ultimate attenuation rate of only 6 dB per octave. A bootstrapping resistor is therefore used to improve performance. Above the cut-off frequency, where the gain of the circuit would otherwise fall off somewhat, R1 has the effect of reinforcing the input signal. Well below the cut off frequency, losses through C4 result in the signal level at Q1 emitter being well below that at the junction of C3 and C4. This results in some of the signal at the junction of C3 and being tapped off through R1, with C3 and R1 effectively forming a second high pass filter network. This eliminates the sluts, initial roll off rate (in fact there is a small and insignificant peak of about 0.5dB above the cut off frequency) and speeds up the attenuation rate to a nominal 12dB per octave.

The low pass filter works in much the same way as the high pass one, except of course, the R and C filter elements have been transposed so as to give the correct filter action.

With the specified component values the rumble filter response falls below unity at approximately 45Hz, reaches the -6 d13 point just above 30Hz, and then falls away at a nominal 12dB per octave. The scratch filter response crosses the unity gain point at about 6k5Hz, reaches the -6dB point at approximately 10kHz, and then falls away at a nominal 12dB per octave.

The amplifier is also reasonably sensitive and will give full output with an input of about 50mV. Input impedance is about 50kR.

The slider from the volume control is connected to the, base of Q1 via a DC blocking capacitor. Q1 is connected as a conventional common emitter amplifier with R2 provides the base bias and R3 acting as the collector load. This stage is directly connected to the second transistor which is a PNP type. In this way the current passing through Q1 provides the bias for the second transistor. The output of the second transistor is connected directly to the speech coil of the loudspeaker. This is not normally good practice since the standing current in the output transistor continually biases the coil either slightly in or out from its usual operating point. However if a large speaker is used, as it should be, this has very little effect and, since we are not aiming at hi-fi, it does not matter.

The tone control comprises C2 and RV2 which are connected between the collector and base of Q1 At high resistance settings RV2 has little effect but on minimum settings the 100nF feeds back the high frequencies out of phase, thus cancelling them.

For this circuit to work properly, R3 must be selected with great care. The value shown here of 39 ohms is a typical one and, although it may be used for initial setting up to ensure the circuit is operating, the value should be found by experiment. If it is too low there will be severe distortion at higher volume settings. If it is too high the current drain will be excessive even though the quality of reproduction will be good.

It is very important that Q2 is fitted with a heat sink as it will get very not.

The speaker impedance is not all that critical and in the prototype speakers with an impedance as low as 8 ohms and as high as 80 ohms all worked well, although changing the speaker impedance will also necessitate a change in the value of R3.

The circuit is basically an op-amp used in the non-inverting amplifier mode. The non-inverting input is biased by R4 and R5 via a decoupling network which is comprised of R3 and C3. C4 and C5 give DC blocking at the input and output respectively. With SW1 open there is virtually 100% negative feedback through R1, R2 and C1, giving the circuit unit gain and a flat response. Closing SW1 brings C2 into circuit, and this de-couples some of the feedback through R1 and R2 at frequencies of more than a few hundred Hz, giving the required rising response. Feedback through C1 at high treble frequencies causes the response to fall away above about 5k5Hz, and prevents the very high frequency harmonics from being excessively emphasized.

The disc is cut at constant amplitude, except from 500Hz to 2120Hz where it is cut at constant velocity. When this disc is replayed with a magnetic pickup, the relative output voltage rises with frequency, due to the fact that the magnetically generated voltage is proportional to the velocity of the stylus as it moves sideways in the groove. To restore the original sound quality, a preamplifier with a frequency response that, gives decreasing output with increasing frequency is required. This response curve is known as the RIAA equalization and it is tailored accurately to fit the cutting and replay processes. The signal level from a magnetic pickup is low, generally 20mVpp and so a low noise pre- amplifier is needed.

The circuit shows a realization of this requirement. The low noise amplifier is the LM381 -made by National Semiconductors. A DC bias control is included (RV1, RV2), and the feedback components generate the RIAA curve. Use screened cable for the wiring to the pickup, keep the circuit away from transformers (and the pickup and its wiring) and connect all the earths together, near to the IC.

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